Quadratic amplitude control circuit for cosite interference cancellation

ABSTRACT

A quadratic amplitude matching system and associated method with an associated tuning control system is provided for continuously and automatically tuning a quadratic amplitude matching filter (QAMF) to a band center of an interfering signal to provide improved rejection of an interfering signal coupled from a transmission antenna into a local receive antenna in the presence of local multi-path, thereby providing improved interference cancellation system performance. The matching control system is provided as an element of an interference cancellation system.

FIELD OF THE INVENTION

The invention relates to the field of radio communication and, inparticular, to the reduction of interference in signals coupled from atransmission antenna into a local receive antenna in the presence of alocal multipath.

Description of the Related Art

Unwanted (i.e., interfering) signals manifest themselves in severalways. Interference can cause a reduction in the sensitivity of areceiver (receiver desensitization), masking of a desired signal,tracking of an undesired interfering signal and loss of the desiredsignal, and processing of the unwanted interfering signal instead of thedesired signal. Each of these manifestations of interference limits thecommunication capabilities of the radio system afflicted by thisproblem. The effects of interference can be some combination of theabsence of usable output from a receiver, false signals from a receiver,and malfunction of a device which is operated by the receiver. Duringemergency situations, the loss and corruption of the desired signal canbe critical.

Unwanted signal interference is generally caused by modulation ofsignals provided to the receiver by the carrier waves, or by thewideband noise, generated by collocated transmitters. Unwanted signalinterference also occurs when frequency-hopping transmitters aretransmitting signals at frequencies that are substantially close to thefrequency of the desired receiver signal (i.e., co-channel operation).Unwanted signal interference can also be caused by “pseudo white-noise”generated by transmitters over a wide band of frequencies on either sideof the transmitter's operating frequency. It is often found incollocated transceiver systems that this “pseudo white-noise” reachesunacceptable levels within the operating band of adjacent receivers.Unwanted signal interference is also attributed to signals (i.e.,spurious emissions) generated by transmitters at odd harmonics of thefundamental frequency of the transmitter output signal. This is causedby the non-linear transfer characteristics of amplifiers in thetransmitter chain, or by passive inter-modulation at the transmitter orreceiver antenna connectors.

In order to substantially reduce and eliminate the undesired interferingsignals while maintaining the spatial benefits afforded by proximatelylocating transceivers, especially frequency-hopping transceivers,several signal processing techniques have been proposed. Thesetechniques include agile filtering, agile filtering with multicouplingand interference cancellation.

When the signal noise and spurious sidebands generated by theinterfering transmitter are strong, broadband, and scenario dependant,standard interference cancellation is inadequate. Changes in thescenario surrounding the platform may vary the coupling between thetransmitter and the protected receiver and thus require adjustment ofsystem parameters in an adaptive process.

Interference cancellation involves sampling the transmitter outputsignal in order to eliminate from the received signal, any interferingsignal having a frequency proximate to the receiver carrier frequency.In co-site environments, a collocated source usually interferes with thereceiver due to the finite isolation between transmit and receiveantennas. This interference in a co-site environment is a combination ofseveral factors, desensitization caused by one or more nearby high-powertransmitter carriers and wideband moderate to low-power interferencecomponents associated with those carriers. These interference componentsare received by the collocated radio and degrade system operation. Thenearby high-power transmitter carrier signals could simply exist as apart of the platform signal environment. Further, the interferingsignals may be classified as either cosite or remote interferers. Acosite interferer is physically collocated with the receiver on aplatform permitting a physical circuit connection from the interferencegenerator to the receiver. A remote interferer is located far enoughfrom the receiver to preclude a physical circuit connection.

A typical Interference cancellation system utilizes a correlation-basedadaptive controller using feedback derived after the cancellationprocess. The system takes a sample of an interference signal and adjuststhe magnitude and phase such that the result is equal in amplitude and180° out of phase with the interference signal at the input of thereceiver. The vector sum of the two signals will cancel, leaving onlythe signal of interest. In practice, however, the two signals are notidentical, due to unwanted distortion in the reference path, as well asdifferences in signal path lengths and non-ideal components in the Tx/Rxsignal paths. Cancellation performance is a function of amplitude andphase match between the interference signal and the sampled signal.Transmission path distortions include time delay, magnitude amplitudeand phase distortion, linear amplitude and phase distortion, andquadratic distortion, correction of each adding a level of performanceenhancement but also adding to system complexity and difficulty inimplementing the corrections.

To suppress a wideband interference signal, the performance of acancellation system is directly proportional to the match between thesampled transmission cancellation signal and the receive pathinterference signal across the signal bandwidth. The interferencecancellation system (ICS) compensates for minor corrections andcomponent drift by controlling a complex weight that implements flatphase and amplitude controls in the adaptive control loop (ACL) tocorrect the magnitude amplitude and phase errors between the two. Thereceive path interference signal provided to the ICS is disrupted bysignal distortions in time of arrival, linear and non-linear (i.e.,quadratic) amplitude, and linear and non-linear phase. The sampledtransmission cancellation signal must be adjusted to match thisdistorted receive path signal as closely as possible to achieve completenulling of the received interference signal. The present disclosureaddresses these concerns by focusing on minimizing mismatch errorscaused by first order non-linear amplitude distortions.

As is well known, cosite interference cancellation requires amplitudeslope matching across the signal bandwidth to achieve a deep null acrossthe band. U.S. Pat. No. 6,693,971 “Wideband co-site interferencereduction apparatus” (Kowalski) issued on Feb. 17, 2004 and assigned toBAE Systems Information and Electronic Systems Integration Inc.(Greenlawn, N.Y.), incorporated by reference herein in its entirety,discloses a method of implementing a near-linear correction of theamplitude slope using an amplitude slope-matching filter. However, adrawback of the system and method of Kowalski is that it also imparts aquadratic shape to the amplitude across the band.

Similarly, the propagation path can also impart a quadratic amplitudemodulation across the band that will be time varying with the changingenvironment. Together, these two distortions limit the nullingperformance of the cosite interference cancellation system.

A need therefore exists for a system and method for continuouslyadjusting the quadratic amplitude of a coupled co-sited transmittersignal before subtracting it from the propagated and received signalwith multipath dispersive distortions to achieve required nulling. Sucha system would also have to be tuned with the transmitter frequency andadjust to changes in the propagation path distortion.

SUMMARY OF THE INVENTION

It is therefore an object of the present disclosure to provide a methodand apparatus for reducing the effects of interference betweencollocated transceivers.

It is an object of the present disclosure to provide a method andapparatus in which proximately located transceivers can simultaneouslytransmit and receive independent signals without substantially affectingthe quality of a desired signal reception.

It is another object of the present disclosure to eliminate the effectsof interference between collocated transceivers utilizing interferencecancellation.

It is a more particular object of the present disclosure to provide amethod and apparatus for providing a quadratic amplitude matchingcapability to an interference cancellation system by implementing aquadratic amplitude matching filter (QAMF).

It is a more particular object of the present disclosure to provide amethod and apparatus for automatically tuning a bank of lobed filters ofthe QAMF such that the signal tracked is near the center of each lobingstructure to generate quadratic shaping structures in the region of atracked signal spectrum.

It is yet another object of the present disclosure to provide a methodand apparatus for tuning this quadratic amplitude matching filter (QAMF)over as large of a band as possible without external interface orcontrol.

The present disclosure provides a quadratic amplitude matching filter(QAMF) architecture and a tuning control system as an element of aninterference cancellation system and associated method for continuouslyand automatically tuning a quadratic amplitude matching filter (QAMF) toa band center of an inserted coupled transmitted signal for improvedinterference cancellation system performance and adjusting to matchpropagation path distortion. More particularly, the QAMF system providesimproved rejection of an interfering signal coupled from a transmissionantenna into a local receive antenna in the presence of local multipath.

The tuning control system and associated method of the presentdisclosure provide improved signal rejection over other possible tuningapproaches by continuously tuning (adjusting) a lobed filter of thetuning control system so that the QAMF has a quiescent flat shape in theregion of the tracked signal spectrum.

In accordance with one embodiment of the present disclosure a tuningcontrol system is provided for reducing interference in signals coupledfrom a transmission antenna into a local receive antenna in the presenceof a local multi-path. The tuning control system interfaces with atime-delay based lobed filter architecture including delay means forforming synchronously locked lobed filters for both a tuning filter fortracking to a predominant interfering signal inserted at an input portand a bank of filters capable of applying a first order quadraticamplitude matching to effect the amplitude shape desired for distortionmatching. The system further includes control means, associated with thedelay means, for tuning the QAMF to track the inserted signal and centerit at the center of the filter, thereby eliminating the need tointerface the control means with the transmitter.

In accordance with one embodiment of the present disclosure, a method isprovided for implementing a first-order quadratic correction to theamplitude of an input signal across its band by the use of a quadraticamplitude matching filter (QAMF), the method comprising: dividing aninput signal into three parallel branches, dynamically adjusting a delaytime (T) in the first branch for tuning a first narrowband RF lobedfilter with one of its quiescent lobes peaked on an interfering signalto be tracked, wherein the first narrowband RF lobed filter is broadenough in bandwidth to implement a first path with near-linear and flatamplitude shape, forming a second, more narrowband RF lobed filter inthe second branch dependent upon the delay time (T) wherein one of thequiescent lobes of the more narrowband RF filter is peaked on theinterfering signal to be tracked but controlled to be more narrow toimplement a second path for downward quadratic amplitude adjustment ofthe inserted signal, forming a simple FIR filter in the third branchdependent upon the delay time, (T), wherein its central form is shapedto form an upward quadratic shape centered on the interfering signal tobe tracked, and wherein the FIR filter upward quadratic area is centeredon the interfering signal to be tracked to implement a third path forupward quadratic amplitude adjustment of the inserted signal, matchingeach of the first, second and third branches to have a uniform pathdelay dependent upon the delay time (T), weighting each of the first,second and third branches according to an external control function, andcombining the first, second and third paths into a single output toallow the QAMF to implement a first-order quadratic amplitude distortionof the input coupled transmitted signal to match the delayed coupledsignal to that of the propagation path for improved interferencecancellation of the inserted signal in an interference cancellationsystem

Also, in accordance with one embodiment of the present disclosure, amethod is provided for continuously and automatically tuning a quadraticamplitude matching filter (QAMF) to a band center for improvedinterference cancellation system performance, the method comprising: a)forming a broadband RF lobed filter having a single quiescent nullwithin a frequency band of interest; b) dynamically adjusting a delaytime (T) for tuning the single quiescent null of the broadband RF lobedfilter to effectively block the inserted signal to be tracked; c)forming a first narrowband RF lobed filter with a quiescent lobe peakcentered on a quiescent null of the inserted signal to be tracked,wherein an output of the first narrowband RF lobed filter output has anear-linear and flat amplitude shape to match a dynamically changingquadratic amplitude distortion of the inserted signal to be tracked; d)forming a second narrowband RF lobed filter with a quiescent lobe peakcentered on the quiescent null of the inserted signal to be tracked,wherein an output of the second narrowband RF lobed filter has adownward quadratic amplitude shape to match a dynamically changingquadratic amplitude distortion of the inserted signal to be tracked; e)forming an FIR filter with a quiescent lobe peak centered on thequiescent null of the inserted signal to be tracked, wherein an outputof the FIR filter has an upward quadratic amplitude shape to match adynamically changing quadratic amplitude distortion of the insertedsignal to be tracked; f) adjusting an in-line path delay of the firstnarrowband RF lobed filter, the second narrowband RF lobed filter andthe FIR filter to have the same throughput delay; and g) adjustingcombining weights of the respective filter outputs of the firstnarrowband RF lobed filter, the second narrowband RF lobed filter andthe FIR filter to implement a corrective quadratic amplitude shaping ofthe inserted signal, thereby matching a dynamically changing quadraticamplitude distortion of the inserted signal to be tracked.

According to one aspect of the method described above, dynamicadjustment of the time-delay element considers both direction and degreein dependence upon the most recent nulling filter output comparisonresult.

In different embodiments, the system may be implemented in discreetcomponents or alternatively as a MMIC. Time delays can be implemented aseither a switched delay or a continuously variable delay through ananalog control voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other objects, features and advantages of the invention willbe apparent from a consideration of the following Detailed DescriptionOf The Invention considered in conjunction with the drawing Figures, inwhich:

FIG. 1 illustrates the general block diagram of an improved cositeinterference cancellation system, according to one embodiment.

FIG. 2 a is a circuit diagram of the general structure of a lobed filterfor use in an improved cosite interference cancellation system,according to one embodiment.

FIG. 2 b illustrates the general structure of the lobed filter, for usein an improved cosite interference cancellation system, according to oneembodiment.

FIGS. 3 a-b are exemplary output waveforms of a lobed filter forillustrating that subtraction, rather than summation, of the output oftwo signal paths of the lobed filter forms an orthogonal filter of thesame repetitive bandwidth as the output of the lobed filter from a basicdelay (T).

FIG. 4 are exemplary resultant output waveforms of a lobed filter formedfrom the summed outputs of a lobed filter having signal pathscharacterized by delays which are odd integer multiples of a basic delay(T), the resultant output waveforms illustrating that a lobe of thesummed outputs is always aligned with a lobe of a lobed filter formedfrom the basic delay (T).

FIG. 5 a-c illustrate three different exemplary tuning scenarios of agenerated wideband RF lobed filter that is orthogonal to the lobe of animaginary (unformed) wideband RF tuning filter.

FIG. 6 is a block diagram of a five-tap finite impulse response (FIR)filter whose pass-band amplitude repeats in the frequency domain,according to one embodiment.

FIG. 7 a is a plot comprising four curves, a first curve representingthe output of a lobed broadband tuning filter, the second curverepresenting a lobed narrowband tuning filter, a third curverepresenting a narrower lobed narrowband filter and a fourth curverepresenting an FIR filter.

FIGS. 7 b & 7 c are plots representing the four curves of FIG. 7 a inincreasingly expanded form.

FIG. 8 is a circuit diagram of a quadratic amplitude matching filter(QAMF) structure where the tuning control comes from an externalcontroller, according to one embodiment.

FIG. 9 is a circuit diagram of an adaptively tuned quadratic control(ATQC) structure where the time delay tuning control circuit has beenintegrated, according to one embodiment.

FIG. 10 is a circuit diagram of the adaptively tuned quadratic control(ATQC) of FIG. 9 incorporated into an interference cancellation systemto improve the cancellation of a local transmitter signal that isreceived in the receive antenna with a time varying modulation due tochanges in local multipath.

FIG. 11 illustrates one embodiment of an improved cosite interferencecancellation system 20 for elimination of interfering signals betweenthree or more co-located transceivers

FIG. 12 illustrates an improved cosite interference cancellation systemfor elimination of interfering signals between a single co-locatedtransceiver and a plurality of receivers to be protected.

DETAILED DESCRIPTION OF THE INVENTION

In the following discussion, numerous specific details are set forth toprovide a thorough understanding of the present invention. However,those skilled in the art will appreciate that the present invention maybe practiced without such specific details. In other instances,well-known elements have been illustrated in schematic or block diagramform in order not to obscure the present invention in unnecessarydetail. Additionally, for the most part, details concerning networkcommunications, electromagnetic signaling techniques, and the like, havebeen omitted inasmuch as such details are not considered necessary toobtain a complete understanding of the present invention and areconsidered to be within the understanding of persons of ordinary skillin the relevant art.

The present description illustrates the principles of the presentdisclosure. It will thus be appreciated that those skilled in the artwill be able to devise various arrangements that, although notexplicitly described or shown herein, embody the principles of thedisclosure and are included within its spirit and scope.

All examples and conditional language recited herein are intended forpedagogical purposes to aid the reader in understanding the principlesof the disclosure and the concepts contributed by the inventor tofurthering the art, and are to be construed as being without limitationto such specifically recited examples and conditions.

Moreover, all statements herein reciting principles, aspects, andembodiments of the disclosure, as well as specific examples thereof, areintended to encompass both structural and functional equivalentsthereof. Additionally, it is intended that such equivalents include bothcurrently known equivalents as well as equivalents developed in thefuture, i.e., any elements developed that perform the same function,regardless of structure.

The functions of the various elements shown in the figures may beprovided through the use of dedicated hardware as well as hardwarecapable of executing software in association with appropriate software.When provided by a processor, the functions may be provided by a singlededicated processor, by a single shared processor, or by a plurality ofindividual processors, some of which may be shared. Moreover, explicituse of the term “processor” or “controller” should not be construed torefer exclusively to hardware capable of executing software, and mayimplicitly include, without limitation, digital signal processor (“DSP”)hardware, read only memory (“ROM”) for storing software, random accessmemory (“RAM”), and nonvolatile storage.

Other hardware, conventional and/or custom, may also be included.Similarly, any switches shown in the figures are conceptual only. Theirfunction may be carried out through the operation of program logic,through dedicated logic, through the interaction of program control anddedicated logic, or even manually, the particular technique beingselectable by the implementer as more specifically understood from thecontext.

Overview

The present disclosure is directed to a tuning control system andassociated method for continuously and automatically tuning a quadraticamplitude matching filter (QAMF) to a band center of a reference inputsignal for improved interference cancellation system performance in acosite interference cancellation system. In some embodiments, the tuningprocess may be performed off-line to preclude the interruption ofprocessing, during an operation stage, with intermediate or finalcontrol signals being transferred to an inline structure to implementthe same control.

The present disclosure provides an automated system and method thatperforms dynamic adjustment of the delay time, tuning a quadraticamplitude matching filter (QAMF) that centers the QAMF filter on thefrequency of its reference input signal as a pre-requisite to adjustingthe QAMF filter for quadratic amplitude control (i.e., amplitudematching). More particularly, the present disclosure provides a novelquadratic amplitude-matching filter (QAMF) to implement dynamic,real-time correction to the quadratic amplitude mismatch. The presentdisclosure further provides a time delay tuning control 202, coupled tothe quadratic amplitude matching filter (QAMF) to provide frequencytuning to the quadratic amplitude matching filter (QAMF) without theneed for an external tuning control signal (e.g., a tuning controlsignal from the transmitter as practiced in the prior art). It should beunderstood, however, that quadratic amplitude control is required, as afurther processing step beyond performing dynamic, real-time tuning. Asis well known, Quadratic amplitude control is performed to adjust thequadratic amplitude-matching filter (QAMF) to the proper weights tomatch the quadratic amplitude distortion of a sampled transmissionsignal to that of the propagation path.

Referring now to the drawings, FIG. 1 is a circuit diagram forillustrating an improved cosite cancellation circuit 20 for eliminatinginterfering signals between radio transmitter 21, as an element oftransceiver 1, and receiver 25, as an element of transceiver 5, wheresystem dynamics cause changes in the coupling between transmit antenna 2and receive antenna 4, co-located on a platform, according to oneembodiment.

It should be understood that each of the transceivers 1, 5 functionindependent of the other such that they alternate in being viewed aseither the interfering transmitter or protected receiver depending uponthe specific needs of the user. However, the system description willonly address a single functional aspect for ease of explanation. Thetransceivers 1, 5 can operate at any RF frequency including, forexample, in the high frequency (HF), very high frequency (VHF) andultra-high frequency (UHF) spectrums.

The improved cosite cancellation circuit 20 for the elimination ofinterfering signals between radio transceivers 1, 5, is adapted to becoupled to transceiver 5, in the illustrative embodiment, or other typeof device, known or envisioned, capable of receiving electronic signals.The transceiver 1 operating in the transmission mode produces electronicsignals for transmission via antenna 2 of transceiver 1. Substantiallycontemporaneously to this signal transmission, other electronic signalsare received by antenna 4 and provided to at least transceiver 5operating in the receiving mode. As is known to happen, in addition tothe signals intended to be received by antenna 4, the co-locatedtransmitter 21 also generates noise and distortion signals whichinterfere with the electronic signals received by the antenna 4 that areto be provided to a receiver 5.

In order to substantially eliminate the effect of the interferingsignals generated from transceiver 1, the novel cancellation circuit 20is electrically coupled to transmission signal 40. In a preferred formof the present invention, a directional coupler 7 is operatively coupledto the output port of transmitter 21. The cancellation circuit 20receives a sample of the filtered transmission signal corresponding tothe transmitter 1 to which it is coupled.

Operation

In operation, transmitter 21 transmits RF transmission signal 40 throughantenna 2 which couples spatially 3 either directly or through amultipath environment into a second antenna 4 connected to a receiver 25on the same platform as interfering transmitter 21. This coupled energyinterferes with the reception in the receiver 25 of its desiredreception of a distant transmission. The interfering transmitter 21 thusbecomes a collocated source of interference. It is desired to protectthe receiver 25 from the interfering transmitter 21. The addition of asimple Interference Cancellation System (ICS) consisting of only acoupled adaptive control loop (ACL) 6 can reduce this interference to alimited extent by sampling the transmission signal 7 and feeding it intothe auxiliary port 8 of the ACL 6 while antenna signal 30, includingboth the interfering propagated reference signal and the desired signal,is fed into the reference port 9 of the ACL 6.

In an environment clear of reflective obstacles (e.g., no multi-pathsources present), the spatially coupled signal 3 from antenna 2 toantenna 4 would be received unchanged except for the propagation delayand the quadratic amplitude matching filter (QAMF) 100 would not berequired. However, in a typical multi-path laden environment, thespatially coupled signal 3 is distorted across the band in a number ofways, one of them being an undesirable quadratic amplitude distortionacross the band of interest which is constant in a stable environmentbut varies with a changing multipath environment of a platform inmotion.

Static v. Dynamic Environments

In a static environment, the cable delay, T_(D) 277, between samplepoint 7 and point 8, the input to ACL 6, is ideally adjusted to be thetypical coupling delay through space from source antenna 2 to receiveantenna 4. This delay, T_(D) 277, is implemented to include the delay ofQAMF 100 and any other in-line delays. The next level of correction isthe amplitude slope matching 278 which is ideally adjusted to match theamplitude slope distortion through space from source antenna 2 toreceive antenna 4. These corrections will change with time in a dynamicenvironment but are not the subject of this disclosure. In a dynamicenvironment, as environmental conditions change with time in anunpredictable manner, a variable quadratic amplitude distortion can beaffected upon the propagated signal resulting in an undesirable mismatchbetween the coupled transmission (i.e., the signal coupled via path 7 to8) and the propagated transmission (i.e., the signal coupled via path2-9) limiting the effectiveness of the applied cancellation.

To correct a dynamically changing quadratic amplitude mismatch betweenthe afore-mentioned signals, the present disclosure provides, in oneaspect, a quadratic amplitude matching filter (QAMF) 100, generallyshown in FIG. 1 and illustrated in more detail in FIG. 7, to implement adynamic correction to the amplitude slope mismatch between the delayedcoupled signal 57 and antenna signal 30, including both the interferingpropagated reference signal and the desired signal.

To successfully track and match the distortion introduced by thedispersive propagated interfering propagated reference signal, containedin antenna signal 30, interference cancellation circuit 20 must firstcontinuously and automatically tune a quadratic amplitude matchingfilter (QAMF) 100, to the reference input interfering transmitted signal40 band center. This continuous and automatic tuning process comprises akey feature of the invention. In a preferred embodiment, the tuningprocess is continuously and automatically performed by a local tuningcontrol system (i.e., time delay tuning control 202), as a quiescentstarting point for performing subsequent operations such as quadraticamplitude adjustment.

It should be understood that the present disclosure is primarilydirected to: (1) an architecture that implements a quadratic amplitudecorrection, (2) the tuning of the quadratic amplitude matching filter(QAMF) 100 as a pre-requisite to performing quadratic amplitudeadjustment, and (3) quadratic amplitude adjustment under control of anadaptive amplitude controller 225 (see FIG. 1 a). It should beunderstood that Adaptive amplitude control adjustment 225 uses standardcontrol algorithms and processes, which are well known in the art, andperipheral to the teachings of the present disclosure. However, Adaptiveamplitude control adjustment is briefly discussed as follows.

Adaptive Amplitude Adjustment

As is well known, RF spectral amplitude adjustment may be implemented byforming a filter of desired shape. Filters of differing shape can beformed in parallel and a controller can select the best match for theapplication but there is often no way of knowing a priori which filterwill best match the application. Another way of selecting a filteroutput, or even generating a new filter from a composite of a number offilters, is to weight and sum each of the filter outputs in a variableweighting structure. A controller is provided which has a feedbackmechanism such that it can change the weighting and summation networkweight values and then evaluate the change. Adaptive amplitude control225 implements this process by monitoring the protected output 58 of ACL6 (See FIG. 1 a) while dithering control lines that adjust the weightsof the quadratic amplitude matching filter (QAMF) under a sequencedetermined by its algorithm and loop feedback.

Referring now to FIG. 2 a, a circuit structure is shown for forming atunable variable lobed filter 250, according to one embodiment. In thisembodiment, the tunable variable lobed filter 250 is implemented using apower divider 252, a variable delay 254 and a summing junction 256. Thetunable variable lobed filter 250 is tunable by changing the variabledelay value [T] 254. Tuning the variable delay value [T] 254 causesexpansion and contraction of each lobe from zero and thus a shift ofevery lobe, beyond the first, up or down in frequency.

FIG. 2 b is an alternative circuit structure 260 of the tunable variablelobed filter 250 implemented with a difference hybrid 258 as asubstitute for the summing junction 256. This creates a functionallysimilar tunable variable lobed filter 250 as described above but hasorthogonal lobes to the structure of FIG. 2 a, providing an importantmathematical relationship to be used in control of the tuning process,as discussed immediately below and also further below with reference toFIG. 3.

The inventor has recognized two important mathematical relationshipsthat together allow tuning over a large bandwidth and control of a morenarrowband filter to provide the desired amplitude shaping effect. Thefirst important mathematical relationship relates to the orthogonalnature of the sine and cosine function of two RF filters simultaneouslyformed from the same power divider 252 and time delay structure whencombined in either a sum or difference port of the tunable variablelobed filter 250, as briefly discussed above. The first recognizedmathematical relationship allows the use of a null at one frequency in asine filter to align with the lobe of the cosine filter, or vice versa,and can be used as a sensitive tuning control, as illustrated in FIG. 3,and described below.

The second important mathematical relationship is the recognition thattwo RF filters, one tuned with time delay T and the other tuned with afurther time delay (2n+1)T, where n is an integer, will always havelobes co-aligned at the center of the wider band lobe. It is noted thatthe relationship is one of the further time delay being an odd multipleof a basic delay T. The implications of such a relationship aredescribed in more detail further below with respect to FIG. 4.

Referring now to FIGS. 3 a-3 b, there is shown an output of a lobedfilter, such as, for example, the tunable variable lobed filter 250 ofFIGS. 2 a and 2 b. The output is represented as curve 390 in FIG. 3 a(and further illustrated in expanded form in FIG. 3 b).

Referring to FIG. 3 a, the output 390 of tunable variable lobed filter250 is shown as a magnitude (cosine) function of the delay difference inthe two paths, i.e., path A and path B, shown in FIG. 2 a. The lobedfilter amplitude of output curve 390 of FIG. 3 a repeats at a regularspacing of BW_(n) equal to (2T)⁻¹. As stated above, in an alternateembodiment, a difference hybrid 258 (See FIG. 2 b) can be substitutedfor the summing junction 256 (See FIG. 2 a) of the tunable variablelobed filter 250 FIG. 2 a. In such an embodiment, the output 390 of thetunable variable lobed filter 250 follows a magnitude (sine) function,represented as curve 391 in FIG. 3 a. Thus, a time delay can be selectedto have the tunable variable lobed filter 250, 260 extend beyond a bandof interest and a corresponding orthogonal filter will have a nullwithin the tuning bandwidth. For example, by extending tunable variablelobed filter 250 of FIG. 2 a beyond a band of interest it will have anull 390 within the tuning band of interest. As a further example, byextending tunable variable lobed filter 260 of FIG. 2 b beyond a band ofinterest, it will have a null 391 within the tuning band of interest.

It should be appreciated that the null to null bandwidth, BW_(n) of thelobed tunable variable lobed filter 250 is inversely proportional to thetime delay, T 254, as shown in FIGS. 2 a and 2 b. Therefore, an increasein the time delay T 254 reduces the bandwidth BW_(n) of the tunablevariable lobed filter 250. Further, by changing the time delay to be anodd multiple of a basic delay T, for example, by (2n+1)T, where n aninteger, the original tunable variable lobed filter 250 is effectivelysplit into (2n+1) lobes. As this always results in an odd number oflobes, one lobe 402, necessarily is always centered with the tuning lobe404 of a broadband tuning filter, as shown in FIG. 4. This singlecentered lobe 402 becomes useful in the quadratic amplitude matchingfilter (QAMF) structure 100 (See FIGS. 7 and 8) to be weighted by theadaptive amplitude control 225 controller to shape the coupled signal tomatch the propagated interfering propagated reference signal, containedin antenna signal 30 in an interference cancellation system. The valueof n used to effectively split the output of the filter into 2n+1 lobescan be adjusted to achieve the desired flatness at quiescent withminimal propagation path distortion as required for slaved lobe filterstructure-flat 103 but can also be adjusted to the value m to providethe required quadratic down shaping required for slaved lobe filterstructure-down 104. It is also contemplated to use the values of n and mas variables for finer tuning control in future envisionedimplementations of the improved cosite interference cancellation system.

FIGS. 5 a-5 c illustrate, by way of example, plots of differentexemplary tuning scenarios to further illustrate the concept ofgenerating a lobed filter orthogonal to the lobe of a broadband tuningfilter. It should be understood that, in accordance with inventionprinciples, a tuning filter lobe of the broadband tuning filter is notnecessary for actual operation, and is not necessarily formed in actualoperation. It will therefore be referred to hereafter as a so-calledimaginary tuning filter lobe. It should be understood, however that thequadratic amplitude matching filter (QAMF) will track the center of theso-called imaginary tuning filter lobe by use of the orthogonal nullformed off-line in the timing delay tuning control (TDTC) 202, as shownin FIGS. 1, 9 and 10. Herein, inline refers to an action or process thatgenerates an immediate change, upon signals passing through, at theoutput of the circuit where offline refers to action or processes thatmay use samples of signals passing through but do not impact the signalspassing through until a result is reached and a change is made to theinline processes.

Each of the plots of FIGS. 5 a-5 c illustrates a common insertion signal511 to be tracked. The insertion signal represents the sample oftransmitted signal 40 to be matched to an undesirable multipath signalreceived in antenna signal 30 to be cancelled by the improved cositecancellation circuit 20 of FIG. 1.

Referring first to FIG. 5 a, four output filter curves are shown 511,512 N_(O), 513 N_(L), 514 N_(H). Output filter curves 512 N_(O), 513N_(L) and 514 N_(H) represent three different filters tuned with aso-called imaginary tuning filter lobe but orthogonal to the imaginarytuning filter lobe such that nulls of the orthogonal filter are alignedwith the peak of a lobe of the original filter formed by the same delay,T. A first filter output curve 512 N_(O) represents the null portion ofa lobed filter, N_(O), formed by current value of delay T, orthogonal tothe tuning filter tuned on frequency with the imaginary tuning filter byusing the same delay T used to form the tuning filter. Using the samedelay used to form both the first filter output curve 512 N_(O) and theimaginary tuning filter, results in a null of the first filter outputcurve 512 N_(O) aligned with the imaginary tuning lobe of the tuningfilter, as shown in FIG. 3.

A second filter output curve 513 N_(L) represents the null portion of alobed filter, N_(L), formed by delay T+ΔT, an incremental step of delaytime T 254 of the circuit of FIG. 2 tuned low in frequency with a pathdelay difference of T+ΔT and results in a null below, or lower than thecurrent center frequency of the imaginary tuning lobe of the broadbandlobed filter.

A third filter output curve 514 N_(H) represents the null of the lobedfilter, N_(H), is tuned high in frequency with a path delay differenceof T−ΔT and results in a null above, or higher than, the current centerfrequency of the imaginary tuning lobe of the broadband lobed filter.

With continued reference to FIG. 5 a, there is shown the condition inwhich the filter, N_(O), is centered at a frequency that is below thefrequency of the insertion signal 511. In this case, the filter N_(H)allows more of the inserted signal energy of the inserted signal 511through, than the filter N_(L) thus providing feedback to theinterference cancellation system to move the tuning filter higher infrequency by decreasing the delay, T.

FIG. 5 b illustrates the case in which the filter, N_(O), is centered ata frequency that is above the frequency of the insertion signal 511. Inthis case, the filter N_(L) allows more of the inserted signal energy ofthe inserted signal 511 through, than the filter N_(H) thus providingfeedback to the interference cancellation system to move the tuningfilter lower in frequency by increasing the delay, T.

FIG. 5 c illustrates the case where conditions when the filter, N_(O),is centered on the inserted signal. In this case, the low and highfilters, N_(H) and N_(L), will pass equal amounts of the inserted signalenergy, thus providing no feedback to change frequency by changing thedelay, T. This state represents a point of stability in tuning suchthat, as the null of the orthogonal filter is aligned with the insertedsignal and thus aligned with the peak of the center of the lobe of theimaginary tuning filter, the inserted signal is thus aligned with thepeak of the quadratic amplitude matching filter (QAMF) center.

It should be understood that the direction of the null shifts as afunction of the time delay introduced by the interference cancellationsystem is inherent to lobed filters which are comprised of a pluralityof nulls originating at zero Hz and repeating at a regular spacing of(2T)⁻¹. Thus an increase in delay T reduces the effective BW_(n),thereby compressing the lobing and shifts the current null to the left,i.e., lower in frequency.

Referring again to FIG. 5 a, the center null 512 N_(O) is representativeof a filter output null which is orthogonal to the corresponding filteroutput formed by the summation of the output of filter signal paths withpath delay differences formed by the inline delay T.

The left null 513 N_(L) represents the null of a filter output having apath delay T+Δt, the output exhibiting a slightly more narrow lobedstructure than the output of a filter signal path having a path delay T,and thus the repetitive lobing shifts to the left, lower in frequency,moving the null below the nominal location using delay T.

The right null 514 N_(H) represents the null of a filter output having apath delay T−Δt, the output exhibiting a slightly wider lobed structurethan the output of a filter signal path having a path delay T, and thusthe repetitive lobing shifts to the right, higher in frequency, movingthe null above the nominal location using delay T.

It should be appreciated that these two filter output curves 513 N_(L),514 N_(H) advantageously allow different amounts of the incident signalenergy to pass through them. In this manner, measurement of the energyfrom the respective filter outputs provides information on a correctivedirection in frequency of the tuning lobe orthogonal filter required forproper tuning.

With continued reference to FIG. 5 a, this figure further illustrates aset of undesirable image nulls 515. It is appreciated that theseundesirable image nulls 515 are a limitation to the tuning bandwidth ofthe tuning control system. They arise by using too large of a value ofdelay T, resulting in an excess of narrow lobes for tuning. It thereforefollows that it is desirable to have as large a tuning bandwidth aspossible to preclude the creation of these image nulls. It is preferredthat tuning to the low edge of the frequency tuning band cannot allowimage nulls to approach the high band limit for inserted signal, or viceversa, or the system may shift lobes of the tuning filter upon a jump intransmitted signal frequency, and cause significant change in subsequentfilter bandwidths and thus shaping amplitude factors. The narrowbandfilter cannot be used for tuning because of this limitation. This showsthe importance of the recognition of the lobe alignment for filtersformed by T and (2n+1)T delays so that the tuning filter lobe can bevery broad for a broad tuning range but still be used to focus a muchmore narrow lobe for quadratic amplitude matching filter (QAMF)function.

As stated above, a primary objective of the tuning control system of thepresent disclosure is to continuously and automatically tune a quadraticamplitude matching filter (QAMF) to an interferer band center as aquiescent starting point for performing quadratic amplitude controladjustment. While it is understood that amplitude control adjustment isnot central to the teachings of the present disclosure, it is understoodthat it is implemented by controlling the weights of the tuned quadraticamplitude matching filter (QAMF), tuned in accordance with inventionprinciples.

The tunable variable lobed filter 250 structure cannot provide thenecessary shaping required for the slaved lobe filter structure-up 105of the quadratic amplitude matching filter (QAMF). A finite impulseresponse filter with structure shown in FIG. 6 can provide the shaperequired while maintaining linear phase.

Referring now to FIGS. 7 a-7 c, there is shown, by way of example, aplot of four curves. For ease of explanation, each signal has beenoffset in level for clarity and each successive plot is an expansion ofthe center area of the previous plot, as indicated by the commoncenterline.

The first curve 610 is representative of an imaginary broadband tuningfilter formed by a delay interval T, corresponding to an off-line lobedtuning filter with an orthogonal null to allow it to track an incomingsignal of interest. The second curve 612 is representative of a lobedfilter tracking the tuning filter with a delay interval (2n+1)T, where nis some integer multiplier of T. In this case, the lobed filter tracksthe off-line broadband tuning filter null, as is true of the first curve610 formed with a delay T, however, in the present case, the filter ismore narrow in bandwidth although still nearly flat in the region of thebandwidth of signal of interest, as generated by Slaved lobe filterstructure-flat 103 and output at 112 (See FIGS. 1 and 8).

The third curve 614 is representative of the lobed filter tracking thetuning filter with a multiplier value of m=28 in the present examplesuch that it tracks the imaginary broadband tuning filter lobe centerand the slaved lobe filter structure-flat 103 (See FIGS. 1 and 8). Thislobe structure is even more narrow in bandwidth than the lobe structuregenerated by slaved lobe filter structure-flat 103 and output at 112(See FIGS. 1 and 8). The presently described lobe structure hasamplitude shaped as a down quadratic in the region of the bandwidth ofsignal of interest, i.e., the “signal bandwidth” region, as generated byslaved lobe filter structure-down 104 and output at 118.

The fourth curve is representative of a more complex FIR filter (e.g.,formed using 5 taps by way of example and not limitation). In thisembodiment, weights having values of 1.0, 1.0, −1.0, 1.0, 1.0 are usedto create an upward quadratic curve in the region of the signal ofinterest as generated by slaved lobe filter structure-up 106 and outputat 122. Further, the tuning filter is tracked using a tap spacing of Twith multiplier of o (o=14 in this example) such that it tracks thetuning filter lobe center, the slaved lobe filter structure-flat 103 andthe slaved lobe filter structure-down 104.

Referring now to FIG. 7 c, there is shown a region labeled “SignalBandwidth” for illustrating the alternate signal path amplitude filtershapings, before weighting and combining, implemented upon the coupledtransmitted signal 40 (see FIG. 1 a) in the quadratic amplitude matchingfilter (QAMF) 100. The coupled transmission signal 40 is desirablyshaped by a cosite cancellation circuit 20 of the protected receiver 25for the purpose of matching the distortion introduced in the propagatedinterfering propagated reference signal, contained in antenna signal 30for improved interference cancellation.

FIG. 8 is a more detailed circuit diagram of the quadratic amplitudematching filter (QAMF) 100 of FIG. 1. In the presently describedembodiment, a quadratic amplitude adjustment 100 is implemented as ablock of three parallel filters 103, 104, 105, each respectively formedin standard finite impulse response filters having differentcharacteristics of amplitude shapings across the band of interest andeach being formed based upon different odd integer multiples of a basicdelay interval T which tunes such structures to a central band ofinterest of an interfering signal.

The three parallel filter blocks include the slaved lobe filterstructure-flat block 103, the slaved lobe filter structure-down block104, and the slaved lobe filter structure-up block 105. Each filterblock 103, 104, 105 uses a common digital control signal W_(T) 129 as atuning signal to track a center frequency with different relativebandwidths, to be described as follows.

Slaved lobe filter structure-flat 103 includes an equalizing delay blockand a simple filter. The signal enters a time delay t_(a) 111 controlledby the W_(T) 129 but internal circuitry is designed to adjust thesetting of the delay implemented in t_(a) 111 to cause the total delaythrough the slaved lobe filter structure-flat 103 to that matching thesimultaneous delays through slaved lobe filter structure-down 104 andslaved lobe filter structure-up 105. The delayed signal enters thesimple filter at a power divider 108 forming two paths, one of whichfeeds directly into one port of a summing junction 110 while the secondpath enters a controlled delay line 109 which is set to a delay (2n+1)Tby the same control W_(T) 129 before entering a second port of thesumming junction 110. This delay corresponds to the delay that wouldtune a similar lobed filter used for amplitude slope control to the bandof interest and bandwidth such that it is nearly flat in the region ofthe signal of interest. The signal exits the summing junction and entersa weighting device

Slaved lobe filter structure-down block 104 is the same structure asslaved lobe filter structure-flat block 103 except that the filter timeconstant for Slaved lobe filter structure-down block 104 is increased tonarrow the lobing of the filter to generate a quadratic shape in thearea of the signal of interest.

Slaved lobe filter structure-down block 104 includes an equalizing delayblock 117 and a simple filter. The signal enters a time delay t_(b) 117controlled by the same W_(T) 129. Internal circuitry (not shown) isincluded in slaved lobe filter structure-down 104 to adjust the settingof the delay implemented in t_(b) 117 to cause the total delay throughthe slaved lobe filter structure-down block 104 to match thesimultaneous delays through slaved lobe filter structure-flat block 103and slaved lobe filter structure-up block 105. The delayed signal entersthe simple filter at a power divider 114 forming two paths, one of whichfeeds directly into one port of a summing junction 116 while the secondpath enters a controlled delay line 115 which is set to a delay (2m+1)Tby the control W_(T) 129 before entering a second port of the summingjunction 116. This delay is based upon the same tuning interval T buthas a multiplier of (2m+1). Multiplier m establishes the relativebandwidth of the quadratic filter shaping.

Slaved lobe filter structure-up block 105 includes an equalizing delayblock 121 and a simple FIR filter. The signal enters a time delay t_(c)121 controlled by the same W_(T) 129 but internal circuitry is designedto adjust the setting of the delay implemented in t_(c) 121 to cause thetotal delay through the slaved lobe filter structure-up block 105 tothat matching the simultaneous delays through slaved lobe filterstructure-flat block 103 and slaved lobe filter structure-down block104. The delayed signal enters the finite impulse response (FIR) filter120 of multiple taps spaced at intervals based upon the same tuninginterval T but having a multiplier of (2o+1) and individually weightedto generate the desired upward quadratic function at the frequency ofthe signal of interest. The multiplier o will cause the filter functionto track the signal of interest as the frequency changes and the circuitis tuned in response.

The outputs of the three slaved lobe filter shaping structures 103, 104,105 are each respectively weighted and summed to generate the amplitudeshaping necessary to match a received signal which has been distorted ina dispersive multipath propagation path from a co-located transmissionantenna 2. Quiescent values of W_(p1) 113, W_(p2) 119, and W_(p3) 123will be (1.0, 0.0, 0.0), providing minimal shaping are modifiedthereafter by the adaptive amplitude control 225.

FIG. 9 provides the circuit structure of a tuning control system,according to one embodiment, for continuously and automatically tuning aquadratic amplitude matching filter (QAMF) 100 to a band center of areference input signal for improved interference cancellation systemperformance in a cosite interference cancellation system.

An Adaptively Tuned Quadratic Control (ATQC) module 200 comprises twomain elements; an inline quadratic amplitude matching filter (QAMF) 100and an offline Time Delay Tuning Control (TDTC) element 202. Thequadratic amplitude matching filter (QAMF) 100 further includes avariable lobe filter structure (VLFS) 201, slaved lobe filterstructure-down 104 and a slaved lobe filter structure-up 105. It isnoted that Variable lobe filter structure (VLFS) 201 is a variation ofslaved lobe filter structure-flat 103 of FIG. 1 with the time delaystructure split into two blocks to provide a tuning feed.

The variable lobe filter structure (VLFS) 201 implements thefunctionality of a tuned and quiescent amplitude slope matched filtersof the prior art but is constructed in a novel manner as a variation ofa conventional in-line Lobe Filter Structure such that the delay lineforming the lobed filter is split into two blocks of controlled variabletime delay. Specifically, the delay line is split into a first block 205with delay T and a second block 208 with delay 2nT, yielding a totaldelay of (2n+1)T. The first block 205 is used for broadband tuning andthe second block 208 is implemented as a multiple of the first block205, thus making the nearly-flat path more narrowband. As discussedabove, a resulting nearly-flat path filter lobe formed by the (2n+1)Trelationship is centered in the lobe of an imaginary filter orthogonalto the null of the broadband tuning filter lobe formed by the delay Tbut is never actually formed. This establishes one path of the quadraticamplitude matching filter (QAMF) 100. The other two paths are slaved tothe tuning value T and thus track the inserted coupled transmittedsignal. The weights on the three paths are then adjusted for quadraticamplitude matching, as taught above and then fed into port 8 of the ICSfor improved interference cancellation, advantageously requiring nocontrol signals from the transmitter.

The Timing Delay and Tuning Control (TDTC) module 202 uses signalsamples output from the variable lobe filter structure (VLFS) 201 tocontrol the first block, T 205 for tuning, which is central to theteachings of the present disclosure, and the second block, 2nT 208, toimplement the lobed filter function-flat used by the interferencecancellation system, which is a pre-requisite for implementing theamplitude slope adjustment to match the propagated path of theinterfering signal.

It should be understood that the tuning filter lobe is referred toherein as imaginary in the sense that it is never actually formed orused in actual operation but is instead discussed herein to provide amore complete understanding of the interrelationships of the variouscontrol signals and the generation of the ASMF filter.

With continued reference to FIG. 9 reference signal 215 and delayedsignal 216 are sampled and fed into a differencing hybrid 221 to form abroadband RF filter, a first broadband RF filter having a filterresponse 223. Thus the offline sine function filter formed using thevariable T 205 has a null orthogonal to the center of the imaginarytuning filter lobe based upon this T, which is never formed.

In accordance with a method for continuously and automatically tuning aquadratic amplitude matching filter (QAMF) 100 to a band center forimproved interference cancellation system performance, the energypassing through the filter is measured by controller 222. The controllercan then dither the digital delay control 232 by a value ΔT, eitherpositive or negative and re-measure the energy passing through thefilter. In this way, the controller can determine direction to skew thetuning filter to achieve a null on the input signal 203. This assumesincreasing a control voltage causes the delay T to be increasedresulting in a narrowing of the filter lobes while decreasing thecontrol voltage causes the delay T to be decreased resulting in abroadening of the filter lobes. Signs can be easily changed for deviceswith opposite control functions. There are a number of search algorithmscommon in the art to perform this control that provide for desiredrejection of noise and timely convergence. These include but are notlimited to random searching, gradient searching, and perturbation usingorthogonal Walsh functions, each with advantages and disadvantages. Theselected control algorithm is not part of this present disclosure.

FIG. 10 is a more detailed circuit diagram of FIG. 1 for illustrating animproved interference cancellation circuit 20 for elimination ofinterfering signals between radio transmitter 21, and receiver 25 wheresystem dynamics cause changes in the coupling between a transmit andreceive antenna on a platform, according to one embodiment.

A time delayed, quadratic amplitude matched sample of transmissionsignal 40 is output from adaptively tuned quadratic control 200 as thedelayed coupled signal 57 and supplied to auxiliary port 8 of ACL 6.Interfering propagated reference signal, contained in antenna signal 30is fed into reference port 9 of ACL 6. A cancellation signal 65 isgenerated by ACL 6 via the processes of autocorrelation 66, integration67, and finally by applying a complex weight 68 of phase and amplitude.The cancellation signal 65 is provided to summing junction 70. It isnoted that when the cancellation signal 65 is injected into summingjunction 70 it has substantially the same amplitude as the interferingpropagated reference signal, contained in antenna reference signal 71,however, the cancellation signal 65 is manipulated so that it is 180°out of phase with the interfering propagated reference signal receivedby antenna 4 and included in antenna signal 30 so as to substantiallycancel the interfering signal. The adaptive amplitude control 225 ofprior art is still required to adjust the quadratic amplitude matchingfilter (QAMF) 100 to the proper weights to match the quadratic amplitudedistortion of the sampled transmission signal to that of the propagationpath. Adaptive amplitude control 225 implements this process bymonitoring ICS protected output 58 while dithering control lines W_(p1)113, W_(p2) 119, and W_(p3) 123 under a sequence determined by itsalgorithm and loop feedback. As a result, the signal remaining on theprotected output 58 is substantially the same as the received antennasignal 30 provided by receiver antenna 4 without the undesiredcontribution from interfering transmitter 1. ACL 6 is configured as aLeast Mean Square (LMS) analog control loop but those familiar with theart will realize that many different algorithms, implemented at RF anddigital, can serve this function.

The use of sum or difference hybrids in the off-line processing andin-line processing may be switched to design the system for a specifictuning band and slope control lobed filter width. This embodiment isjust one configuration.

If the tuning information is available from the transmitter, it could beused for a table lookup of the starting point for the value of T. Thus,when the transmitter switched frequency, tuning would start atapproximately the correct value. These stored values may be a one-timeset value at manufacture or may be updated every time the frequency isvisited.

Referring now to FIG. 11 there is shown four co-located interferingtransmitters 21 a-21 d, by way of example and not limitation. Four areshown for ease of explanation. To counteract the multiple interferingtransmitters 21 a-21 d, and thus reduce or minimize cosite interference,the improved cosite interference cancellation system 20 includes acommon adaptive amplitude control 225 of prior art operably coupled to acommon summing junction for four Interference Cancellation Systems (ICS)6 a-6 d, and four independent Adaptively Tuned Quadratic Control (ATQC)module 200 a-200 d comprised of two main elements; an inline quadraticamplitude matching filter (QAMF) 100 and an offline Time Delay TuningControl (TDTC) element 202 operably coupled to a common summing junctionfor four Interference Cancellation Systems (ICS) 6 a-6 d. Four of whichare shown for ease of explanation and not limitation. In this manner,the cosite interference cancellation process described above withreference to FIG. 10 is independently applied to each interferingtransmitter 21 a-21 d to protect the single receiver 25. This figureshows a preferred embodiment with common, shared antenna signal 30,summing junction 70 and antenna reference signal 71. The function of theVariable Lobe Filter Structure (VLFS) 201 a-201 d are in-line and mustbe independent but the function of the Time Delay Tuning Control (TDTC)element 202 and adaptive slope control 225 can be shared throughmultiplexing techniques implemented in prior art of adaptive arrayswhere the correlation and integration functions were shared. In otherembodiments, the ICS summing junctions are daisy-chained for the use ofa standard building block at the cost of additional potential noiseinsertions and longer convergence times because of signal interaction.

Referring now to FIG. 12 there is shown an improved cosite interferencecancellation system 20 for elimination of interfering signals between asingle co-located transceiver 21 and a plurality of receivers to beprotected. In the presently described embodiment, it is desired toprotect a multiplicity of receivers, 25 a-25 d, four of which are shownby way of example and not limitation. To protect the plurality ofreceivers 25 a-25 d, each receiver is coupled to a correspondingInterference Cancellation Systems (ICS) 6 a-6 d operably coupled withassociated independent Adaptively tuned quadratic control 200 a-200 deach comprises three main elements; a Variable Lobe Filter Structure(VLFS) 201, a Time Delay Tuning Control (TDTC) element 202, and a commonadaptive amplitude control 225 of prior art.

The foregoing is construed as only being an illustrative embodiment ofthis invention. Persons skilled in the art can easily conceive ofalternative arrangements providing a functionality similar to thisembodiment without any deviation from the fundamental principles or thescope of the invention.

1. A method for continuously and automatically tuning a quadraticamplitude matching filter (QAMF) to an inserted signal to allow trackingof the inserted signal to match a dynamically changing quadraticamplitude distortion of the inserted signal for improved interferencecancellation system performance, the method comprising: a) forming animaginary broadband RF lobed filter having a single quiescent nullwithin a frequency band of interest; b) dynamically adjusting a delaytime (T) for tuning the single quiescent null of the imaginary broadbandRF lobed filter to effectively block the inserted signal to be tracked;c) forming a first narrowband RF lobed filter with a quiescent lobe peakcentered on a quiescent null of the inserted signal to be tracked,wherein an output of the first narrowband RF lobed filter output has anear-linear and flat amplitude shape to match a dynamically changingquadratic amplitude distortion of the inserted signal to be tracked; d)forming a second narrowband RF lobed filter with a quiescent lobe peakcentered on the quiescent null of the inserted signal to be tracked,wherein an output of the second narrowband RF lobed filter has adownward quadratic amplitude shape to match a dynamically changingquadratic amplitude distortion of the inserted signal to be tracked; e)forming an FIR filter with a quiescent lobe peak centered on thequiescent null of the inserted signal to be tracked, wherein an outputof the FIR filter has an upward quadratic amplitude shape to match adynamically changing quadratic amplitude distortion of the insertedsignal to be tracked; f) adjusting an in-line path delay of the firstnarrowband RF lobed filter, the second narrowband RF lobed filter andthe FIR filter to have the same throughput delay; and g) adjustingcombining weights of the respective filter outputs of the firstnarrowband RF lobed filter, the second narrowband RF lobed filter andthe FIR filter to implement a corrective quadratic amplitude shaping ofthe inserted signal, thereby matching a dynamically changing quadraticamplitude distortion of the inserted signal to be tracked.
 2. The methodof claim 1, wherein the respective outputs of the first and secondnarrowband RF lobed filters and the FIR filter are dependent upon thedynamically adjusted delay time (T) for tuning the single quiescent nullof the imaginary broadband RF lobed filter.
 3. The method of claim 1,wherein the first and second narrowband RF lobed filters and the FIRfilter are orthogonal to the imaginary broadband RF lobed filter in aquiescent state.
 4. The method of claim 1, wherein said step (g) ofadjusting combining weights of respective filter outputs of the firstnarrowband RF lobed filter, the second narrowband RF lobed filter andthe FIR filter parallel filters is performed under control of anexternal amplitude control signal.
 5. The method of claim 1, wherein thefirst and second narrowband RF lobed filters and the FIR filter arecomprised of a plurality of lobes formed within the frequency span of asingle lobe of the imaginary broadband RF lobed tuning filter.
 6. Themethod of claim 5, wherein one of the plurality of lobes of thenarrowband RF lobed filter is peaked on the inserted signal to betracked.
 7. The method of claim 6, wherein said one of said plurality oflobes peaked on the inserted signal to be tracked is a quiescent lobe ofan amplitude sloped matching filter (ASMF) function.
 8. The method ofclaim 6, wherein said one of said plurality of lobes is peaked on theinterfering signal to be tracked and is centered at a null of thebroadband RF lobed filter.
 9. The method of claim 5, wherein one of theplurality of lobes of the imaginary broadband RF lobed tuning filter iscentered on the inserted signal to be tracked has an amplitude shape tocorrect a dynamically changing quadratic amplitude distortion of theinserted signal to be tracked.
 10. The method according to claim 1,wherein the broadband RF lobed filter has a null to null bandwidthsubstantially twice the desired tuning bandwidth.
 11. The methodaccording to claim 1, wherein said step (b) of dynamically adjusting adelay time is performed via a control process.
 12. The method accordingto claim 11, wherein the control process comprises: a) measuring an RFenergy output from the broadband RF filter; b) applying a deltaincrement to the delay time (T); c) re-measuring the RF energy outputfrom the broadband RF filter; d) comparing the measured and there-measured values of RF energy; and e) determining if the applied deltaincrement to the delay time (T) improves the tuning as measured by anenergy output from the broadband RF filter; f) if improved, updating thedelay time (T) by adding a positive delta increment as T=T+ΔT; g) ifunimproved, updating the delta delay time (ΔT) by changing sign of deltadelay time increment as ΔT=−ΔT.
 13. The method of claim 1, whereinincreasing the control voltage causes the delay T to be increasedresulting in a narrowing of the filter lobes.
 14. The method of claim13, wherein narrowing the filter lobes results in a downward frequencyshift of the individual lobes and nulls of the filter.
 15. The method ofclaim 1, wherein decreasing the control voltage causes the delay T to bedecreased resulting in a widening of the filter lobes.
 16. The method ofclaim 15, wherein widening the filter lobes results in an upwardfrequency shift of the individual lobes and nulls of the filter.
 17. Aninterference cancellation system, comprising: (A) an adaptively tunedquadratic control (ATQC) module (200) for providing a tuning proceduredirected to a quadratic amplitude matching filter (QAMF) (100) controlfunction to the band of interference and for externally controlling theQAMF control function subsequent to said tuning, said adaptively tunedquadratic control (ATQC) module (200) comprising: a) an inline variablelobe filter structure (VLFS) (201) for providing controlled variabletime delay for generating a broadband RF tuning filter formed by a delayT and a first narrowband inline control filter for providing a near-flatquadratic control filter lobe formed by a first delay (T) and a seconddelay (2nT) yielding a total time delay of (2n+1)T centered in the nullof the broadband RF tuning filter and weighted by an external amplitudecontrol signal; b) a second narrowband inline control filter as a slavedlobe filter structure-down (104) for providing a down quadraticamplitude control filter lobe formed by a delay of (2m+1)T centered inthe null of the RF tuning filter and weighted by an external amplitudecontrol signal; c) a third narrowband inline control filter as a slavedlobe filter structure-up (105) for providing a up quadratic amplitudecontrol filter lobe formed by a simple FIR filter with inter-tap spacingdelay of (2o+1)T centered in the null of the tuning filter and weightedby an external amplitude control signal; d) an offline time delay tuningcontrol (TDTC) element (202) for receiving signal samples output fromthe inline variable lobe filter structure (VLFS) 201 to control a firstvariable time delay element (T) 205 of the inline variable lobe filterstructure (VLFS) 201 to provide said controlled variable time delay toform said broadband RF tuning filter and a second variable time delayelement (2nT) (208) of the inline variable lobe filter structure (VLFS)201 to provide said controlled variable time delay to generate saidslope control filter lobe. (B) an adaptive control loop (6) foradjusting a complex weighting of the delayed coupled signal (57) tomaximally cancel a propagated reference signal.
 18. The interferencecancellation system of claim 17, wherein the inline variable lobe filterstructure (VLFS) 201 comprises: a) said first variable time delay line(T) (205) for providing broadband tuning of an imaginary tuning filterlobe; and b) a second variable time delay line (2nT) (208) for providingmore narrowband tuning of the imaginary tuning filter lobe relative tosaid first delay element; wherein said first variable time delay element(205) and second variable time delay element (2nT) (208) yield a totaltime delay of (2n+1)T centered in the null of the tuning filter andskewed by an external slope control signal.
 19. The interferencecancellation system of claim 17, wherein the second variable time delayline (2nT) 208 is an integer multiple of said first time delay line (T)205.
 20. The interference cancellation system of claim 17, wherein theadaptive control loop (6) comprises: a reference port (9) for receivingthe an antenna signal (30); an auxiliary port (8) for receiving adelayed and matched coupled signal (57); a complex correlator (66) forgenerating error correlation signal (72) an integrator (67) to smoothtransients on the error correlation signal (72) to form the adaptiveweight control signals (73); a complex phase and amplitude weightingdevice (68) having a first input and a second input, said first inputreceiving said delayed and matched coupled signal (57), said secondinput receiving a complex adaptive weight control signal (73) to weightthe delayed and matched coupled signal (57) to produce a weighteddelayed coupled signal (65); and forming a weighted delayed coupledsignal (65) as a mirror image of a propagated reference signal,contained in antenna signal 30; a summing junction (70) having a firstand second input, said first input for receiving said weighted delayedcoupled signal (65) output from said complex phase and amplitudeweighting device (68), said second input for receiving the antennareference signal (71) to yield a protected output signal (58).
 21. Theinterference cancellation system of claim 17 wherein said propagatedreference signal comprises at least a transmission signal (40)propagated through an uncontrolled path from a first antenna (2) andreceived at a second antenna (4).
 22. The interference cancellationsystem of claim 20, wherein forming the weighted delayed coupled signal(65) as a mirror image of the transmitted reference signal indicatesthat it is equal in amplitude and 180° out of phase with a portion ofthe transmitted signal (40) in the antenna reference signal (71). 23.The interference cancellation system of claim 17, wherein said antennasignal (30) includes both said propagated reference signal and at leastone other signal.
 24. The interference cancellation system of claim 23,wherein the at least one other signal is a desired signal anticipated bya protected receiver (25).
 25. A method for implementing a first-orderquadratic correction to the amplitude of an input signal across its bandby the use of a quadratic amplitude matching filter (QAMF), the methodcomprising: a) dividing an input signal into three parallel signalpaths; b) dynamically adjusting a delay time (T) in a first signal pathfrom among said three parallel signal paths for tuning a firstnarrowband RF lobed filter with one of its quiescent lobes peaked on aninterfering signal to be tracked; c) forming a second more narrowband RFlobed filter in a second signal path from among said three signal pathsdependent upon the delay time (T) wherein one of the quiescent lobes ofthe second narrowband RF filter is peaked on the interfering signal tobe tracked; d) forming a simple FIR filter in the third branch dependentupon the delay time, (T), having a filter configuration in the form ofan upward quadratic shape centered on the interfering signal to betracked, e) matching each of the first, second and third signal paths tohave a uniform path delay dependent upon the delay time (T); f)weighting each of the first, second and third signal paths according toan external control function, and g) combining the first, second andthird signal paths into a single output to allow facilitate theimplementation of a first-order quadratic amplitude distortion of theinput signal via the QAMF to match the delayed coupled signal to that ofthe propagation path for improved interference cancellation of theinserted signal in an interference cancellation system.
 26. The methodof claim 25, wherein the first narrowband RF lobed filter issufficiently broad enough in bandwidth to implement a first pathcharacterized by a near-linear and flat amplitude shape.
 27. The methodof claim 25, wherein the second more narrowband RF lobed filter iscontrolled to be more narrow than the first narrowband RF lobed filterto implement said second signal path for downward quadratic amplitudeadjustment of the inserted signal.
 28. The method of claim 25, whereinthe FIR filter upward quadratic area is centered on the interferingsignal to be tracked to implement said third signal path for upwardquadratic amplitude adjustment of the inserted signal.